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9077262.ppt

  1. 1. Chapter 6 Static CMOS Circuits Boonchuay Supmonchai Integrated Design Application Research (IDAR) Laboratory August , 2004; Revised - June 28, 2005
  2. 2. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 2 Goals of This Chapter  In-depth discussion of CMOS logic families  Static and Dynamic  Pass-Transistor  Nonratioed and Ratioed Logic  Optimizing gate metrics  Area, Speed, Energy or Robustness  High Performance circuit-design techniques
  3. 3. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 3 Combinational Sequential Output = f(In) Combinational Logic Circuit Out In Combinational Logic Circuit Out In State Output = f(In, Previous In) Combinational vs. Sequential Logic
  4. 4. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 4 Static CMOS Circuits  At every point in time (except during the switching transients) each gate output is connected to either VDD or VSS via a low-resistance path.  The outputs of the gates assume at all times the value of the Boolean function, implemented by the circuit (ignoring, once again, the transient effects during switching periods)  This is in contrast to the dynamic circuit class, which relies on temporary storage of signal values on the capacitance of high impedance circuit nodes
  5. 5. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 5 PUN and PDN are dual logic networks F(In1,In2,…InN) VDD In1 PUN PDN In2 InN In1 In2 InN Static Complementary CMOS PMOS transistors only NMOS transistors only Pull-Up Network: make a connection from VDD to F when F(In1,In2,…InN) = 1 Pull-Down Network: make a connection from F to GND when F(In1,In2,…InN) = 0
  6. 6. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 6 Threshold Drops PDN PUN CL VDD  VDD 0  CL VDD VDD VDD  CL 0  VDD CL S D S D S D VGS S D VGS VDD VDD - VTn 0 |VTp|
  7. 7. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 7 A B A B Construction of PDN  Transistors can be thought as a switch controlled by its gate signal  NMOS switch closes when switch control input is high NMOS Transistors pass a “strong” 0 but a “weak” 1 A • B Series = NAND A + B Parallel = NOR
  8. 8. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 8 Construction of PUN  PMOS switch closes when switch control input is low. PMOS Transistors pass a “strong” 0 but a “weak” 1 A B A B Series = NOR A • B = A + B Parallel = NAND A + B = A • B
  9. 9. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 9 Duality of PUN and PDN  PUN and PDN are dual networks  De Morgan’s theorems  A parallel connection of transistors in the PUN corresponds to a series connection of the PDN  Complementary gate is naturally inverting (NAND, NOR, AOI, OAI)  Number of transistors for an N-input logic gate is 2N A + B = A • B [!(A + B) = !A • !B or !(A | B) = !A & !B] A • B = A + B [!(A • B) = !A + !B or !(A & B) = !A | !B]
  10. 10. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 10 A • B A B A B A B F 0 0 1 0 1 1 1 0 1 1 1 0 Example: CMOS NAND gate VDD A B F PDN: G = A · B PUN: F = A + B Conduction to GND Conduction to VDD G(In1, In2, …, InN) = F(In1, In2, …, InN)
  11. 11. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 11 Example: CMOS NOR gate A + B A B A B VDD A B F 0 0 1 0 1 0 1 0 0 1 1 0 A B F
  12. 12. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 12 D A B C D A B C OUT = D + A • (B + C) Complex CMOS Gate Derive PUN hierarchically by identifying sub-nets
  13. 13. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 13 Cell Design: An Introduction  Standard Cells  A general purpose logic  Synthesizable  Same height but varying width  Datapath Cells  For regular, structured designs (arithmetic)  Including some wiring in the cell  Fixed height and width
  14. 14. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 14 Example of A Standard Cell Cell boundary N Well Cell height 12 metal tracks Metal track is approx. 3 + 3 Pitch = repetitive distance between objects 2 Rails ~10 In Out VDD GND Cell height is “12 pitch” Minimum-Size Inverter
  15. 15. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 15 signals Routing channel VDD GND Standard Cell Layout Methodology – 1980s Routing channel What logic function is this?
  16. 16. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 16 M2 No Routing channels VDD GND M3 VDD GND Mirrored Cell Mirrored Cell Standard Cell Layout Methodology – 1990s
  17. 17. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 17 Contains no dimensions Represents relative positions of transistors Inverter In Out VDD GND NAND2 A Out VDD GND B Stick Diagrams
  18. 18. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 18 OAI21 Logic Graph C A B B A C i j j VDD X X i GND A B C PUN PDN A B C X = C • (A + B) Node of the circuit Transition Control
  19. 19. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 19 Two Stick Diagrams of C • (A + B) A B C X VDD GND X C A B VDD GND Crossover can be eliminated by re-ordering inputs
  20. 20. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 20 j VDD X X i GND A B C A B C  For a single poly strip for every input signal, the Euler paths in the PUN and PDN must be consistent (the same) Consistent Euler Path  An uninterrupted diffusion strip is possible only if there exists an Euler path in the logic graph Euler path: a path through all nodes in the graph such that each edge is visited once and only once.
  21. 21. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 21 A B C D C A B B A D C D X = (A+B)•(C+D) VDD X X GND A B C PUN PDN D OAI22 Logic Graph
  22. 22. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 22 B A D VDD GND C X  Some functions have no consistent Euler path like x = !(a + bc + de) (but x = !(bc + a + de) does!) OAI22 Layout
  23. 23. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 23 A B A  B XNOR XOR A B A  B A  B A B A B A  B  How many transistor in each?  Can you create the stick diagrams for the lower left circuit? XNOR/XOR Implementation
  24. 24. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 24 VGS2 = VA –VDS1 VGS1 = VB M1 M2 M3 M4 A B F= A • B A B Cint D D S S 0.5/0.25 NMOS 0.75 /0.25 PMOS weaker PUN VTC Characteristics are dependent upon the data input patterns applied to the gate (so the noise margins are also data dependent!) VTC is Data-Dependent
  25. 25. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 25 Observation I  The difference between the blue and the orange lines results from the state of internal node int between the two NMOS Devices.  The threshold voltage of M2 is higher than M1 due to the body effect (),  VSB of M2 is not zero (when VB = 0) due to the presence of Cint VTn1 = VTn0 and VTn2 = VTn0 + ((|2F| + Vint) - |2F|)
  26. 26. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 26 Review: CMOS Inverter - Dynamic tpHL = f(Rn, CL) tpHL = 0.69 Reqn CL tpHL = 0.69 (3/4 (CL VDD)/ IDSATn ) = 0.52 CL / (W/Ln k’n VDSATn ) VDD Rn Vout Vin = V DD CL propagation delay is determined by the time to charge and discharge the load capacitor CL
  27. 27. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 27 Review: Designing for Performance  Reduce CL  Increase W/L ratio of the transistor  the most powerful and effective performance optimization tool  watch out for self-loading!  Increase VDD  only minimal improvement in performance at the cost of increased energy dissipation  Slope engineering - keeping signal rise and fall times smaller than or equal to the gate propagation delays and of approximately equal values  good for performance and power consumption
  28. 28. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 28 A Req A CL A Rn A Rp B Rp B Rn Cint NAND Rp A A Rn CL INVERTER Rp Rp Rn Rn CL Cint B A A B NOR Switch Delay Model
  29. 29. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 29 Input Pattern Effects on Delay  Delay is dependent on the pattern of inputs  Low to high transition  both inputs go low  delay is 0.69 Rp/2 CL since two p-resistors are on in parallel  one input goes low  delay is 0.69 Rp CL  High to low transition  both inputs go high  delay is 0.69 2Rn CL  Adding transistors in series (without sizing) slows down the circuit CL A Rn A Rp B Rp B Rn Cint NAND
  30. 30. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 30 Delay Dependence on Input Patterns A=B=10 A=1, B=10 A=1 0, B=1 time [ps] Voltage [V] Input Data Pattern Delay (psec) A=B=01 67 A=1, B=01 64 A= 01, B=1 61 A=B=10 45 A=1, B=10 80 A= 10, B=1 81 NMOS = 0.5m/0.25 m, PMOS = 0.75m/0.25 m, CL = 100 fF
  31. 31. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 31 Transistor Sizing Basic  Inverter as a reference circuit Device Transconductance (See Supplement 1) kn = kn’(W/L)n = (µnox/tox)((W/L)n kp = kp’(W/L)p = (µpox/tox)((W/L)p For rise time equal to fall time, kp = kn (Rp = Rn) Out In VDD (W/L)p (W/L)n CL Because µp ~ µn /2 (W/L)p = 2 (W/L)n The size of PMOS must be twice as large as that of NMOS 2 1
  32. 32. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 32 Transistor Sizing: NAND and NOR 2 2 2 2 1 1 4 4 CL A Rn A Rp B Rp B Rn Cint NAND Rp Rp Rn Rn CL Cint B A A B NOR Symmetric Response RPUN = RPDN Rp  1/(W/L)p Rn  1/(W/L)n RPDN = Rn + Rn RPUN = Rp + Rp
  33. 33. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 33 Note on Transistor Sizing  By assuming RPUN = RPDN, we ignores the extra diffusion capacitance introduced by widening the transistors.  In DSM, even larger increases in the width are needed due to velocity saturation.  For 2-input NANDs, the NMOS transistors should be made 2.5 times as wide.  NAND implementation is clearly preferred over a NOR implementation, since a PMOS stack series is slower than an NMOS stack due to lower carrier mobility
  34. 34. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 34 1 2 2 2 4 4 8 8 Transistor Sizing: Complex CMOS Gate D A B C D A B C 6 6 12 12  Red sizing assuming RPUN = RPDN  Follow short path first; note PMOS for C and B,4 rather than 3 (average in pull-up chain of three = (4+4+2)/3)  Also note structure of pull-up and pull-down to minimize diffusion cap. at output (e.g., single PMOS drain connected to output)  Green for symmetric response and for performance (where Rp = 3Rn)  Sizing rules of thumb: PMOS = 3 * NMOS
  35. 35. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 35 Fan-In Considerations D C B A D C B A CL C3 C2 C1 Distributed RC model (Elmore delay) tpHL = 0.69 Reqn(C1+2C2+3C3+4CL) Propagation delay deteriorates rapidly as a function of fan-in – quadratically in the worst case.
  36. 36. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 36 Notes on Fan-In Considerations  While output capacitance makes full swing transition from VDD to 0, internal nodes only swing from VDD-VTn to GND  C1, C2, and C3, each includes junction capacitance as well as the gate-to-source and gate-to-drain capacitances (turned into capacitances to ground using the Miller effect)  For W/L of 0.5/0.25 NMOS and 0.375/0.25 PMOS, values are on the order of 0.85 fF  CL = 3.47 fF with NO output load (all of diffusion capacitance = intrinsic capacitance of the gate itself).  tpHL = 85 ps (simulated as 86 ps).  The simulated worst case low-to-high delay was 106 ps.
  37. 37. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 37 tp as a Function of Fan-In Gates with a fan-in greater than 4 should be avoided. quadratic linear tpLH t p (psec) fan-in tpHL tp
  38. 38. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 38 tp as a Function of Fan-Out All gates have the same drive current. tpNOR2 t p (psec) eff. fan-out tpNAND2 tpINV Slope is a function of “driving strength”
  39. 39. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 39 tp as a Function of Fan-In and Fan-Out  Fan-in: quadratic due to increasing resistance and capacitance  Fan-out: each additional fan-out gate adds two gate capacitances to CL tp = a1FI + a2FI2 + a3FO Parallel Chain Serial Chain
  40. 40. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 40 Distributed RC line M1 > M2 > M3 > … > MN (the FET closest to the output should be the smallest) Can reduce delay by more than 20%; decreasing gains as technology shrinks Fast Complex Gates: Design Technique 1  Transistor sizing  as long as fan-out capacitance dominates  Progressive sizing InN CL C3 C2 C1 In1 In2 In3 M1 M2 M3 MN
  41. 41. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 41 Notes on Design Technique 1  With transistor sizing, if the load capacitance is dominated by the intrinsic capacitance of the gate, widening the device only creates a “self loading” effect and the propagation delay is unaffected (and may even become worse).  For progressive sizing, M1 have to carry the discharge current from M2 (C1), M3 (C2), … MN and CL so make it the largest.  MN only has to discharge the current from MN (CL)(no internal capacitances).  While progressive sizing is easy in a schematic, in a real layout it may not pay off due to design-rule considerations that force the designer to push the transistors apart  increasing internal capacitance.
  42. 42. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 42 charged charged delay determined by time to discharge CL, C1 and C2 delay determined by time to discharge CL discharged discharged Fast Complex Gates: Design Technique 2  Input re-ordering  when not all inputs arrive at the same time CL C2 C1 In3 In2 In1 M1 M2 M3 critical path 1 1 01 charged CL C2 C1 In1 In2 In3 M1 M2 M3 critical path 1 01 charged 1 Place latest arriving signal (critical path) closest to the output can result in a speed up.
  43. 43. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 43 Example: Sizing and Ordering Effects D C B A D C B A CL = 100 fF C3 C2 C1 3 3 3 3 4 4 4 4 4 5 6 7 Progressive sizing in pull-down chain gives up to a 23% improvement. Input ordering saves 5% critical path A – 23% critical path D – 17%
  44. 44. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 44 F = ABCDEFGH Fast Complex Gates: Design Technique 3  Alternative logic structures  Reduced fan-in results in deeper logic depth Reduction in fan-in offsets, by far, the extra delay incurred by the NOR gate.
  45. 45. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 45 Notes on Design Technique 3  Reducing fan-in increases logic depth of the circuit  More stages but each stage has smaller delay  Only simulation will tell which of the two alternative configurations is faster and has lower power dissipation.
  46. 46. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 46 CL CL Fast Complex Gates: Design Technique 4  Isolating fan-in from fan-out using buffer insertion  Optimizing the propagation delay of a gate in isolation is misguided. Reduce CL on large fan-in gates, especially for large CL, and size the inverters progressively to handle the CL more effectively
  47. 47. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 47 tpHL = 0.69 (3/4 (CL VDD)/ IDSATn ) = 0.69 (3/4 (CL Vswing)/ IDSATn ) Fast Complex Gates: Design Technique 5  Reducing the voltage swing  linear reduction in delay  also reduces power consumption  But the following gate is much slower!  Or requires the use of “sense amplifiers” on the receiving end to restore the signal level (memory design)
  48. 48. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 48 Sizing Logic Paths for Speed  Frequently, input capacitance of a logic path is constrained  Logic also has to drive some capacitance  Example: ALU load in an Intel’s microprocessor is 0.5pF  How do we size the ALU data path to achieve maximum speed?  We have already solved this for the inverter chain – can we generalize it for any type of logic?
  49. 49. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 49 Inverter Chain: Recap CL In Out 1 2 N 1 f f N-1 For given N: Ci+1/Ci = Ci/Ci-1 = f = (CL/Cin) N  For optimum performance, we try to keep f ~ 4, which give us the number of stages, N.  Can the same approach (logical effort) be used for any combinational circuit?
  50. 50. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 50 Delay of a Complex Logic Gate  For a complex gate, we expand the inverter chain equation  tp0 is the intrinsic delay of an inverter  f is the effective fan-out (Cext/Cg) - also called the electrical effort  p is the ratio of the intrinsic (unloaded) delay of the complex gate and a simple inverter (a function of the gate topology and layout style)  g is the logical effort
  51. 51. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 51 Notes on Delay of a Logic Gate Gate delay: D = h + p Effort delay Intrinsic delay Effort delay: h = g f Logical Effort Electrical Effort (effective fan-out)  Logical effort first defined by Sutherland and Sproull in 1999.  In a simpler format, = Cout/Cin
  52. 52. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 52 Intrinsic Delay Term, p  The more involved the structure of the complex gate, the higher the intrinsic delay compared to an inverter Ignoring second order effects such as internal node capacitances Gate Type p Inverter 1 n-input NAND n n-input NOR n n-way mux 2n XOR, XNOR n 2n-1
  53. 53. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 53 Logical Effort Term, g  Logical effort of a gate, g, presents the ratio of its input capacitance to the inverter capacitance when sized to deliver the same current  g represents the fact that, for a given load, complex gates have to work harder than an inverter to produce a similar (speed) response Gate Type g (for 1 to 4 input gates) 1 2 3 4 Inverter 1 NAND 4/3 5/3 (n+2)/3 NOR 5/3 7/3 (2n+1)/3 Mux 2 2 2 XOR 4 12
  54. 54. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 54 Notes on Logical Effort  Inverter has the smallest logical effort and intrinsic delay of all static CMOS gates  Logical effort of a gate tells how much worse it is at producing an output current than an inverter (how much more input capacitance a gate presents to deliver same output current)  Logical effort is a function of topology, independent of sizing  Logical effort increases with the gate complexity  Electrical effort (Effective fanout) is a function of load/gate size
  55. 55. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 55 A + B A B A B A A A 2 1 Cunit = 3 2 2 2 2 Cunit = 4 4 4 1 1 Cunit = 5 Example of Logical Effort  Assuming a PMOS/NMOS ratio of 2, the input capacitance of a minimum-sized inverter is three times the gate capacitance of a minimum-sized NMOS (Cunit) A • B A B A B
  56. 56. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 56 Delay as a Function of Fan-Out  The slope of the line is the logical effort of the gate  The y-axis intercept is the intrinsic delay  Can adjust the delay by adjusting the effective fan-out (by sizing) or by choosing a gate with a different logical effort normalized delay fan-out f intrinsic delay effort delay Gate Effort: h = fg
  57. 57. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 57 1 a b c CL 5 Path Delay: Complex Logic Gate Network  Total path delay through a combinational logic block tp =  tp,j = tp0 (pj + (fj gj)/ )  So, the minimum delay through the path determines that each stage should bear the same gate effort f1g1 = f2g2 = . . . = fNgN  Consider optimizing the delay through the logic network how do we determine a, b, and c sizes?
  58. 58. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 58 Path Delay Equation Derivation  The path logical effort, G =  gi  And the path effective fan-out (path electrical effort) is F = CL/g1  The branching effort accounts for fan-out to other gates in the network b = (Con-path + Coff-path)/Con-path  The path branching effort is then B =  bi  The total path effort is then H = GFB  So, the minimum delay through the path is N D = tp0 ( pj + (N H)/ )
  59. 59. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 59 Example: Complex Logic Gates  For gate i in the chain, its size is determined by  For this network  F = CL/Cg1 = 5  G = 1 x 5/3 x 5/3 x 1 = 25/9  B = 1 (no branching)  H = GFB = 125/9, so the optimal stage effort is H = 1.93  Fan-out factors are f1=1.93, f2=1.93 x 3/5 = 1.16, f3 = 1.16, f4 = 1.93  So the gate sizes are a = f1g1/g2 = 1.16, b = f1f2g1/g3 = 1.34 and c = f1f2f3g1/g4 = 2.60 4 j=1 i -1 si = (g1 s1)/gi  (fj/bj) 1 a b c CL 5
  60. 60. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 60 Example – 8-input AND
  61. 61. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 61 Summary: Method of Logical Effort  Compute the path effort: F = GBH  Find the best number of stages N ~ log4F  Compute the stage effort f = F1/N  Sketch the path with this number of stages  Work either from either end, find sizes: Cin = Cout*g/f Reference: Sutherland, Sproull, Harris, “Logical Effort, Morgan-Kaufmann 1999.
  62. 62. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 62 Summary: Key Definitions Sutherland, Sproull Harris
  63. 63. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 63 VDD VSS PDN In1 In2 In3 F RL Load Resistive N transistors + Load • VOH = VDD • VOL = RPN RPN + RL • Assymetrical response • Static power consumption • • tpL= 0.69 RLCL Ratioed Logic  N transistors + Load  VOH = VDD  VOL = RPDN / (RPDN + RL)  Asymmetrical Response  Static Power consumption Plow  tpL = 0.69 RLCL Goal: To Reduce the number of devices over complementary CMOS
  64. 64. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 64 VDD VSS In1 In2 In3 F VDD VSS PDN In1 In2 In3 F VSS PDN Depletion Load PMOS Load depletion load NMOS pseudo-NMOS VT < 0 Ratioed Logic: Active Loads VDD VSS In1 In2 In3 F VDD VSS PDN In1 In2 In3 F VSS PDN Depletion Load PMOS Load depletion load NMOS pseudo-NMOS VT < 0
  65. 65. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 65 Pseudo NMOS NAND and NOR D C B A CL F NAND D C B A CL F NOR  Psedo-NMOS is useful when area is most important  Reduce transistor counts  Used occasionally for large fan-in gates
  66. 66. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 66 Pseudo-NMOS Inverter Characteristics  Assumptions:  NMOS resides in linear mode  VOL is small relative to the gate drive (VDD-VT))
  67. 67. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 67 Pseudo-NMOS Inverter VTC 0.0 0.5 1.0 1.5 2.0 2.5 0.0 0.5 1.0 1.5 2.0 2.5 3.0 W/Lp = 4 W/Lp = 2 W/Lp = 1 W/Lp= 0.25 W/Lp = 0.5 V out (V) Vin (V) Size VOL (V) Pstat (µW) tpLH (ps) 4 0.693 564 14 2 0.273 298 56 1 0.133 160 123 0.5 0.064 80 268 0.25 0.031 41 569 NMOS size = 0.5 µm/0.25 µm Larger pull-up device not only improves performance (delay) but also increases power dissipation and lowers noise margins by increasing VOL
  68. 68. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 68  M1 >> M2  The idea is to reduce static power consump- tion by adjusting the load  Load M2 when there are not too many inputs (A, B, C, or D) active  Switch to Load M1 when all inputs are active (thus require high amount of current to drive) Improved Loads: Adaptive Load A B C D F CL M1 M2 M1 >> M2 Enable VDD Adaptive Load Enable
  69. 69. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 69 Improved Loads: DCVSL VDD VSS PDN1 Out VDD VSS PDN2 Out A A B B M1 M2 Differential Cascode Voltage Switch Logic (DCVSL)
  70. 70. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 70 B A A B B B Out Out XOR-NXOR gate DCVSL Example
  71. 71. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 71 DCVSL Characteristics  Dual Rail Logic  Each input is provided in complementary format and each gate produces complementary output  Increasing complexity  Rail-to-Rail Swing  No static power dissipation  Sizing of the PMOS relative to PDN is critical to functionality, not just performance  PDNs must be strong enough to bring outputs below VDD - |VTp|
  72. 72. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 72 DCVSL Transient Response 0 0.2 0.4 0.6 0.8 1.0 -0.5 0.5 1.5 2.5 Time [ns] V olta g e [V] A B A B A,B A,B Transient Response of a 2-input AND/NAND gate. How does it look like? tin->out = 197 ps tin->out = 321 ps
  73. 73. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 73 A B X Y X = Y if A and B X Y A B X = Y if A or B NMOS Transistors in Series/Parallel  Primary inputs drive both gate and source/drain terminals  NMOS switch closes when the gate input is high Remember - NMOS transistors pass a strong 0 but a weak 1
  74. 74. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 74 PMOS Transistors in Series/Parallel  Primary inputs drive both gate and source/drain terminals  PMOS switch closes when the gate input is low Remember - PMOS transistors pass a strong 1 but a weak 0 X = Y if A and B = A + B X = Y if A or B = A  B A B X Y X Y A B
  75. 75. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 76 A 0 B B F= AB 0.5/0.25 0.5/0.25 0.5/0.25 1.5/0.25 B = VDD, A = 0VDD A = VDD, B = 0VDD A = B = 0VDD V out , (V)  Pure PT logic is not regenerative - the signal gradually degrades after passing through a number of PTs (can fix with static CMOS inverter insertion) VTC of PT AND Gate
  76. 76. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 77 Complementary PT Logic (CPL) B A A B PT Network F F B A A B Inverse PT Network F F A A B F=A+B B B B OR/NOR F=A+B A A B F=A·B B B B AND/NAND F=A·B F=AB F=AB A A A A B B XOR/XNOR
  77. 77. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 78 CPL Properties  Differential, so complementary data inputs and outputs are always available (don’t need extra inverters)  Static, since the output defining nodes are always tied to VDD or GND through a low resistance path  Design is modular; all gates use the same topology, only the inputs are permuted.  Simple XOR makes it attractive for structures like adders  Fast! (assuming number of transistors in series is small)  Additional routing overhead for complementary signals  Still have static power dissipation problems
  78. 78. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 79 NMOS-Only PT Driving an Inverter  Threshold voltage drop causes static power consumption (M2 may be weakly conducting forming a path from VDD to GND)  Notice VTn increases of pass transistor due to body effect (VSB) Vx does not pull up to VDD, but VDD – VTn VGS In = VDD A = VDD Vx M1 M2 B S D Out
  79. 79. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 80 Voltage Swing of PT Driving an Inverter  Body effect – large VSB at X - when pulling high (B is tied to GND and S charged up close to VDD)  So the voltage drop is even worse Vx = VDD - (VTn0 + ((|2f| + Vx) - |2f|)) Time (ns) Voltage (V) In Out X = 1.8V In = 0  VDD VDD X Out 0.5/0.25 0.5/0.25 1.5/0.25 D S B
  80. 80. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 81 Cascaded NMOS-Only PTs B = VDD Out M 1 y M2 A = VDD C = VDD x G S G S x M1 B = VDD Out y M2 C = VDD A = VDD = VDD - VTn1 Swing on y = VDD - VTn1 - VTn2 Swing on y = VDD - VTn1  Pass transistor gates should never be cascaded as on the left  Logic on the right suffers from static power dissipation and reduced noise margins
  81. 81. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 82 M1 M2 A Mn x B Out Solution 1: Level Restorer  For correct operation Mr must be sized correctly (ratioed)  Full swing on x (due to Level Restorer) so no static power consumption by inverter  No static backward current path through Level Restorer and PT since Restorer is only active when A is high Level Restorer Mr A B X Out Mr 0 0VDD 0 VDD OFF VDD 0VDD VDD 0 ON
  82. 82. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 83 Transient Level Restorer Circuit Response Voltage (V) Time (ps) W/Lr=1.75/0.25 W/Lr=1.50/0.25 W/Lr=1.25/0.25 W/Lr=1.0/0.25 node x never goes below VM of inverter so output never switches  Restorer has speed and power impacts:  Increases the capacitance at x, slowing down the gate  Increases tr (but decreases tf) W/Ln=0.50/0.25, W/L1=0.50/0.25, W/L2=1.50/0.25
  83. 83. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 84 Notes on Level Restorer  Pull down must be stronger than restorer (pull up) to switch node X  If resistance of restorer transistor is too small (too wide transistor) it is impossible to bring the voltage at node X below the switching threshold of the inverter, and the inverter never switches!  Sizing of Mr is critical for DC functionality, not just performance!!  It belongs to Dynamic Logic Family
  84. 84. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 85 Solution 2: Multiple VT Transistors  Technology solution: Use (near) zero VT devices for the NMOS PTs to eliminate most of the threshold drop (body effect still in force preventing full swing to VDD) Out In2 = 0V In1 = 2.5V A = 2.5V B = 0V low VT transistors sneak path on off but leaking Watch out for subthreshold current flowing through PTs
  85. 85. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 86 Solution 3: Transmission Gates (TGs)  Full swing bidirectional switch controlled by the gate signal C, A = B if C = 1 A B C C B C = VDD C = GND A = VDD B C = VDD C = GND A = GND  Most widely used solution A B C C
  86. 86. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 87 Resistance of TG Resistance (k) Rp Rn Req Rp Rn 2.5V 0V 2.5V Vout W/Ln=0.50/0.25 W/Lp=0.50/0.25  TG is not an ideal switch - series resistance  Req is relatively constant ( about 8kohms in this case), so can assume has a constant resistance Req = Rp || Rn
  87. 87. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 88 S S S In2 In1 F F = !(In1  S + In2  S) GND VDD In1 In2 S S S S F TG Multiplexer
  88. 88. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 89 off off B A A  B Transmission Gate XOR  F always has a connection to VDD or GND - not dynamic  No voltage drop 6 Transistors
  89. 89. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 90 Transmission Gate XOR B A F = A‘ VDD (B) 0 (B’)  When B = 1, the circuit behaves as if it is an inverter, hence F(B = 1) = A’ off off 1
  90. 90. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 91 Transmission Gate XOR  When B = 0, the circuit acts as a transmission gate, hence F(B = 0) = A (TG ensures no voltage drop) B A F = A when A = 0 weak 0 weak 1 when A = 1 VDD (B’) 0 (B) on on 0
  91. 91. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 92 Transmission Gate XOR B A A‘ B + A B’  Combine the results using Shannon’s expansion theorem, F = B·F(B = 1) + B’·F(B = 0) = A’B+AB’ = A  B
  92. 92. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 93 TG Full Adder Sum Cout A B Cin • 16 Transistors, no more than 2 PTs in series • Full swing • Similar delay for Sum and Carry
  93. 93. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 95 Delay of a TG Chain C C C C VN V1 Vi Vi+1 5 0 5 0 5 0 5 0 Vin  Delay of the RC chain (N TG’s in series) is tp(Vn) = 0.69 kCReq = 0.69 CReq (N(N+1))/2  0.35 CReqN2 Req Req Req Req Vin C C C C V1 Vi Vi+1 VN
  94. 94. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 96 Notes on TG Chain Delay  Delay grows quadratically in N (in this case in the number of TGs in series) and increases rapidly with the number of switches in the chain.  E.g., for 16 cascaded minimum-sized TG’s, each with an Req of 8kohms.  The node capacitance is the sum of the capacitances of two NMOS and PMOS devices (junctions and drains).  Capacitance values is approx. 3.6 fF for low to high transitions.  The delay through the chain is  tp = 0.69 CReq(N(N+1))/2 = 0.69 x 3.6fF x 8kΩ x (16x17)/2 = 2.7ns
  95. 95. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 97  Delay of buffered chain (M TG’s between buffer) tp = 0.69 N/M CReq (M(M+1))/2 + (N/M - 1) tpbuf Mopt = 1.7  (tpbuf/CReq )  3 or 4 TG Delay Optimization  Can speed it up by inserting buffers every M switches Vin VN M C 5 0 5 0 5 0 C C 5 0 5 0 5 0 C C C
  96. 96. B.Supmonchai 2102-545 Digital ICs Static CMOS Circuits 98 Notes on Delay Optimization  Buffered chain is now linear in N  Quadratic in M but M should be small  This buffer insertion technique works to speed up the delay down long wires as well.  Consider 16TG chain example. Buffers = inverters (making sure correct polarity is output).  For 0.5micron/0.25micron NMOSs and PMOSs in the TGs,  simulated delay with 2TG per buffer is 154 ps,  for 3TGs is 154ps, and for 4TG is 164ps.  The insertion of buffering inverters reduces the delay by a factor of almost 2.

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