1. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN
INTERNATIONAL JOURNAL OF ELECTRONICS AND
0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME
COMMUNICATION ENGINEERING & TECHNOLOGY (IJECET)
ISSN 0976 – 6464(Print)
ISSN 0976 – 6472(Online)
Volume 4, Issue 2, March – April, 2013, pp. 144-150
IJECET
© IAEME: www.iaeme.com/ijecet.asp
Journal Impact Factor (2013): 5.8896 (Calculated by GISI) ©IAEME
www.jifactor.com
PREDICTION OF VIABLE PERFORMANCE OF WIRELESS SENSOR
NETWORK BY USING FINITE-DIFFERENCE TIME-DOMAIN
S.R.Shankar a, Dr.G.Kalivarathanb
a
Research Scholar, CMJ University, Meghalaya, Shillong.
b
Principal/ PSN Institute of Technology and Science, Tirunelveli, Tamilnadu, Supervisor,
CMJ University, Shillong. Email:sakthi_eswar@yahoo.com
ABSTRACT
Wireless Sensor Networks (WSNs) offer a promising solution to monitor the physical
world around us. A WSN is comprised of a large number of sensing devices, often referred to
as motes or sensor nodes, which are deployed at the region of interest. The wireless sensor
nodes have processing and communication capabilities, which enable them to autonomously
gather information from the environment and then to generate and deliver “report-messages”
to the remote base stations (remote users). The economic benefits of WSNs are mainly due to
the exclusion of expensive infrastructure required by the wired sensor networks. The Finite-
Difference Time-Domain (FDTD) method introduced powerful tool for solving various
electromagnetic (EM) problems. The development of a 3-D FDTD method for planar devices
is presented in this work. This method offers an accurate design technique for new type of
microstrip filters. A signal estimation technique was developed in order to reduce the FDTD
computation time. By using this signal estimation technique, the number of FDTD iterations
was reduced up to five times. A design algorithm uses FDTD and Neural Networks. This is
much faster than the FDTD method alone. Mobile communications systems require preselect
filters with enhanced properties. This work presents the research on novel microstrip filters.
The technology required by the newly developed filters is economical, no short-circuit
elements and no lumped components are needed. The designs can be easily extended for
planar HTS technology. An emphasis is put on the development of dual mode filters and
filters with cross-coupled novel resonators. This introductive section presents the work on
low-pass and band-pass conventional filters.
Keywords: Finite-Difference Time-Domain, Mobile communications, Resonator, Microstrip,
Dual mode resonators
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2. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN
0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME
1.0 INTRODUCTION
The FDTD signal excitation contains a voltage source below the microstrip line. The
signal needs to propagate a certain distance along the line to let the transient modes to vanish
and reach their true modal nature. In order to minimize the computational domain, the pulse
propagation along a simple input microstrip line is first simulated. After the signal acquires
the correct transversal profile, the pulse is copied at the input of the microstrip device in order
to be analyzed. The same input signal can be used in several simulations of various planar
structures with the substrate and the FDTD grid not changed. FDTD method allows the
analysis of the electromagnetic field in the planar structure at different time instances. As
expected, the FDTD simulations expose the concentration of the electromagnetic energy just
underneath the microstrip lines and patches. The analysis in time domain, easily illustrates
the incident, reflected and transmitted signals. For example, the plot of the electric field
component Ez the propagation of a pulse along a bent microstrip line. After 550 time steps,
the incident pulse is still on the input line. After additional 400 time steps (∆t =0.27 ps), the
pulse turned left, and propagated along a wider microstrip.
2.0 DISPERSION EFFECTS
The fundamental propagation mode for the microstrip line is considered as
approximating the Transversal Electric and Magnetic (TEM) mode, when the fields are
oscillating only in a perpendicular plane on the direction of propagation. For an ideal TEM
mode, not including the material effects, the pulse should not encounter any dispersion. In
practice, the microstrip properties are considered frequency independent for a frequency
bandwidth up to 2 GHz, when designing on low (εr=2.55) dielectric constant substrate.
However, the dispersion effect increases with the increase of the dielectric constant and the
working frequency. The FDTD simulations clearly illustrate the dispersion effects in time
domain. The Ez field just underneath a straight microstrip line versus time at different
locations. The distances y from the source plane to the measuring points are given in ∆y units.
In this case, the substrate was alumina (Al2O3) having a dielectric constant εr=8.88 and
thickness h=0.535 mm. The 50 line width was w50=0.546 mm. The FDTD grid had
∆x=∆y=0.5636 mm and ∆z=0.127 mm. The time step ∆t=0.27 ps and the incident Gaussian
pulse had T=28 ∆t in width and T0=4 T as initial delay. While at the source position, the
pulse is an undistorted Gaussian, after propagating a certain distance, the amplitude decreases
and the pulse broadens with a strong negative tail. Since the dielectric layers were considered
non-dispersive and the numerical dispersion of FDTD is negligible, the pulse distortion
observed along a microstrip line is intrinsic to the fundamental “Quasi-TEM” propagating
mode.
3.0 FDTD ANALYSIS OF DIFFERENT TYPES OF MICROSTRIP DEVICES
Several microstrip devices have been analyzed to verify accuracy of the developed
FDTD method. The method was applied to numerous microstrip devices manufactured on
substrates having different values for dielectric constant and thickness. In some simple cases,
the simulated scattering S parameters were compared with the S parameters provided by
commercial software. In other cases, the simulated response was compared with
measurements or data taken from literature. A linear low impedance resonator on a substrate
a dielectric constant of εr=2.38 and a thickness of h=0.71 mm was simulated using the
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3. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN
0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME
developed FDTD method. The 50 line had the width of w50=1.87 mm. The resonator line
width was wreson=3.07 and the resonator length was lreson=15.54 mm. For such a simple
device, a commercial software, such as Touchstone, modeled the step discontinuities and
could provide accurate S parameters.
4.0 GAP END COUPLED LINEAR RESONATOR
An end coupled linear resonator presents a higher overall quality factor (Q-factor)
than the linear low impedance section discussed above. Consequently, the output signal
decays slowly and the FDTD simulation requires a large number of time steps to complete
the simulation. The gap end coupled linear resonator was designed as having the line width
wreson=w50= 0.61 mm, the length lreson=10 mm and the coupling gap s=0.3 mm. The used
substrate had dielectric constant of εr=9.98 and thickness of h=0.635 mm. The used FDTD
grid had ∆x=∆y=0.1525 mm and ∆z=0.127 mm and the time step was ∆t=0.27 ps.
5.0 DUAL MODE RESONATORS AND FILTERS
Dual mode resonators (DMR) are resonators perturbed in such a way that two
resonating modes, initially degenerate, can couple each other. A DMR offers a dual mode
filter (DMF) behavior, when certain conditions on the input and output couplings are
satisfied. A planar design of dual mode filters was presented using λ meander resonators. A
resonator forms a 2-pole filter and consists of a meander loop with the input and output
structures, two optional stubs for independent tuning of the resonant frequencies of the
orthogonal modes, and a stub providing a coupling between modes. Two types of filter are
possible depending on the stubs location. For the symmetric filter, the stub is located on the
AA' plane in Fig. 4.5 and the frequency response has two transmission zeros located on both
sides of the passband. The asymmetric filter has the stub on BB' plane and does not present
any transmission zero. Four pole elliptic filters, also exhibit transmission zeros, but, unlike in
the 2-pole symmetric filter, the position of the zeros can be fully controlled. Each of these
two rings has three lines attached: an input (or output) line, a line providing major coupling
between rings, and a line providing minor coupling. Moreover, each ring has a stub for tuning
of the center frequency of one of the modes and obviously a stub providing coupling between
orthogonal modes. The input/output structure can be realized in many ways, but for
optimized sensitivity they have been carried out as sections of coupled transmission lines.
Coupling between rings have been realized by using capacitive gaps. The lines between rings
and coupling elements provide appropriate transformations from the gaps or in/out structures
as well as a spatial separation between rings.
6.0 QUASI FRACTAL DUAL MODE RESONATORS
The DMF patch filters have good power handling properties but they occupy a larger
surface area. In order to reduce the patch size, a technique from microstrip antenna design
was borrowed. When slots are cut in the square patch, The perturbation required for the dual
mode effect is provided by the slots’ asymmetry. The coupling between the modes can be
controlled by the difference in the length of the diagonal slots. For the tuning configuration in
Fig. 4.26a, the insertion loss decreased to approximately 12 dB. The filter presents two
transmission zeros on each side of the pass-band
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4. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN
0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME
7.0 RESULTS
The dispersion effect increases with the increase of the dielectric constant and the
working frequency. The FDTD simulations clearly illustrate the dispersion effects in time
domain. The dispersive effects increase with frequency, therefore these absorbing boundary
conditions gave spurious numerical reflections at higher frequency.
Figure1. The pulse is guided by the microstrip line along Oy axis
Figure2. The pulse changes direction after a mittered corner and reaches
Wider microstrip
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5. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN
0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME
The accuracy and the consistency of the present developed FDTD method were proved by
comparing the results of a simulation on a meander line to the measurements. The performed
numerical experiments showed that a straight transition from the lumped source to the
microstrip line provides a better matching than using a tapered line.
Figure3. Calculated S21 (dotted line) and S11 (continuous line), and measured, S11 (*) for a
meander line on Al2O3 with εr=9.98, h=0.635 mm, w = 0.61 mm
The decimation of y2 with the desampling rate desra=200 corresponded to an increase in the
time step. The very small FDTD time step was required by the stability Courant criterion, but
it could cause a coarse frequency step, when translated in frequency domain. The signal y3
resulted from decimation was provided to the ARMA algorithm. The obtained ARMA
coefficients could be considered as the coefficients of an Infinite Impulse Response (IIR)
filter, which could identify the estimated signal. In this case, the order of the IIR filter was K
= M = 16.
Figure4. S parameters (in dB) versus frequency (GHz) of the meander loop
Dual mode filter
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6. International Journal of Electronics and Communication Engineering & Technology (IJECET), ISSN
0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME
8.0 CONCLUSION
Dual mode filters (DMF) can offer a solution when a narrow band is desired. Open
loop and patch DMF are investigated and DMF for GSM / GPRS bands are designed. A
novel quasi-fractal resonator is developed in order to reduce the square DMF patch size.
Filters with cross-coupled resonators are investigated due to their ability to show generalized
Chebyshev (or quasi-elliptic) response. Filters with cross-coupled open half-wavelength
loops are designed. For mobile communications bands, the simply half-wavelength resonators
are inconveniently long, therefore novel type of resonators are developed. Original theoretical
principles of size reducing for these resonators are presented. The newly developed filters
take up to 32% of the surface of a simple square half-wavelength resonator, both being
designed for 900 MHz. The coupling coefficients of the novel resonators, function of their
relative positions are obtained using the 3D-FDTD method. The external quality factor
functions of the input / output line positions are obtained in the same way. An iterative
ARMA signal estimation technique was developed in order to reduce the FDTD computation
time. This is of importance especially in the case of the analysis of narrowband resonating
structures. Therefore, with the present technique, the required number of iterations can be
reduced up to five times, keeping the same accuracy of the results. Finally, a new design
technique using FDTD method and Neural Networks was developed and applied to a
microstrip filter. The total design time was reduced twofold. The ARMA signal estimation
technique was first utilized to reduce the computation time for each FDTD run. Secondly, the
number of FDTD simulations was decreased using the device model provided by a neural
network with the ARMA coefficients at the output. The trained network was then
incorporated in an optimization procedure for a microstrip filter design.
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0976 – 6464(Print), ISSN 0976 – 6472(Online) Volume 4, Issue 2, March – April (2013), © IAEME
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