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cmos vlsi design

cmos vlsi design

- 1. ELEC516/10 Lecture 9 1 ELEC 516 VLSI System Design and Design Automation Spring 2010 Lecture 9 - Low Power Digital CMOS Design Reading Assignment: Rabaey: Chapter 4 and 7,11 Note: some of the figures in this slide set are adapted from the slide set of “ Digital Integrated Circuits” by Rabaey, Copyright UCB 2002
- 2. ELEC516/10 Lecture 9 2 Why worry about power? -- Heat Dissipation Besides low power required for portable applications !!!! Power is a very big concern in today’s advanced technologies. Power produces heat on the chip which has to be carried off through the chip socket expensive packaging solutions
- 3. ELEC516/10 Lecture 9 3 Motivation for low power design • Packaging costs • Power supply rail design • Chip and system cooling costs • Noise immunity and system reliability • Battery life (in portable systems) • Environmental concerns – ICT equipment accounted for 10% of total US commercial energy usage in 2010 and may reach 20% by 2020 – Energy Star compliant systems
- 4. ELEC516/10 Lecture 9 4 Why worry about power — Portability Multimedia Terminals Laptop Computers Digital Cellular Telephony BATTERY (40+ lbs) Year Nominal Capacity (Watt-hours / lb) Nickel-Cadium Ni-Metal Hydride 65 70 75 80 85 90 95 0 10 20 30 40 50 Rechargable Lithium Expected Battery Lifetime increase over next 5 years: 30-40%
- 5. ELEC516/10 Lecture 9 5 Why worry about power? -- Standby Power Drain leakage will increase as VT decreases to maintain noise margins and meet frequency demands, leading to excessive battery draining standby power consumption. 8KW 1.7KW 400W 88W 12W 0% 10% 20% 30% 40% 50% 2000 2002 2004 2006 2008 Standby Power Source: Borkar, De Intel Year 2002 2005 2008 2011 2014 Power supply Vdd (V) 1.5 1.2 0.9 0.7 0.6 Threshold VT (V) 0.4 0.4 0.35 0.3 0.25 …and phones leaky!
- 6. ELEC516/10 Lecture 9 6 Power and Energy Figures of Merit • Power consumption in Watts – determines battery life in hours • Peak power – determines power ground wiring designs – sets packaging limits – impacts signal noise margin and reliability analysis • Energy efficiency in Joules – rate at which power is consumed over time • Energy = power * delay – Joules = Watts * seconds – lower energy number means less power to perform a computation at the same frequency
- 7. ELEC516/10 Lecture 9 7 Power versus Energy Watts time Power is height of curve Watts time Approach 1 Approach 2 Approach 2 Approach 1 Energy is area under curve Lower power design could simply be slower Two approaches require the same energy
- 8. ELEC516/10 Lecture 9 8 PDP and EDP • Power-delay product (PDP) = Pav * tp = (CLVDD 2)/2 – PDP is the average energy consumed per switching event (Watts * sec = Joule) – lower power design could simply be a slower design l allows one to understand tradeoffs better 0 5 10 15 0.5 1 1.5 2 2.5 Vdd (V) Energy-Delay (normalized) energy-delay energy delay Energy-delay product (EDP) = PDP * tp = Pav * tp 2 l EDP is the average energy consumed multiplied by the computation time required l takes into account that one can trade increased delay for lower energy/operation (e.g., via supply voltage scaling that increases delay, but decreases energy consumption)
- 9. ELEC516/10 Lecture 9 9 • Dynamic power – Charge-discharge current: • Dominant source of power (CV2 per transition) – Short circuit current (both NMOS and PMOS on during transit) • <10% of c/d current if transitions are fast • Subthreshold leakage (transistors not OFF completely) – Becoming important 10-30% active power in <0.18um techn – Diode leakage from reverse source and drain diodes (neglig) – Gate leakage (no longer negligible due to very thin gate oxide) Where Does Power Go in CMOS?
- 10. ELEC516/10 Lecture 9 10 Dynamic Power Consumption Vin Vout CL Energy/transition = C L * V dd 2 Power = Energy/transition * f = C L * V dd 2 * f Need to reduce C L , V dd, and f to reduce power. Vdd Not a function of transistor sizes!
- 11. ELEC516/10 Lecture 9 11 Dynamic Power Consumption - Revisited Power = Energy/transition * transition rate = CL * V dd 2 * f 0 1 = C L * V dd 2 * P 0 1* f = C EFF * V dd 2 * f Power Dissipation is Data Dependent Function of Switching Activity CEFF = Effective Capacitance = C L * P0 1
- 12. ELEC516/10 Lecture 9 12 Power Consumption is Data Dependent Example: Static 2 Input NOR Gate Assume: P(A=1) = 1/2 P(B=1) = 1/2 P(Out=1) = 1/4 P(01) = 3/4X 1/4 = 3/16 Then: = P(Out=0).P(Out=1) CEFF = 3/16 * CL
- 13. ELEC516/10 Lecture 9 13 Transition Probabilities for Basic Gates
- 14. ELEC516/10 Lecture 9 14 Transition Probability of 2-input NOR Gate
- 15. ELEC516/10 Lecture 9 16 Inter-signal Correlations B A Z X P(Z=1) = P(B=1) & P(A=1 | B=1) 0.5 0.5 (1-0.5)(1-0.5)x(1-(1-0.5)(1-0.5)) = 3/16 (1- 3/16 x 0.5) x (3/16 x 0.5) = 0.085 Reconvergent • Determining switching activity is complicated by the fact that signals exhibit correlation in space and time – reconvergent fan-out • Have to use conditional probabilities
- 16. ELEC516/10 Lecture 9 17 Logic Restructuring Chain implementation has a lower overall switching activity than the tree implementation for random inputs Ignores glitching effects Logic restructuring: changing the topology of a logic network to reduce transitions A B C D F A B C D Z F W X Y 0.5 0.5 (1-0.25)*0.25 = 3/16 0.5 0.5 0.5 0.5 0.5 0.5 7/64 15/256 3/16 3/16 15/256 AND: P01 = P0 x P1 = (1 - PAPB) x PAPB
- 17. ELEC516/10 Lecture 9 19 Input Ordering Beneficial to postpone the introduction of signals with a high transition rate (signals with signal probability close to 0.5) A B C X F 0.5 0.2 0.1 B C A X F 0.2 0.1 0.5 (1-0.5x0.2)x(0.5x0.2)=0.09 (1-0.2x0.1)x(0.2x0.1)=0.0196
- 18. ELEC516/10 Lecture 9 20 How about Dynamic Circuits? Power is Only Dissipated when Out=0! Mp Me VDD PDN f In1 In2 In3 Out f CEFF = P(Out=0).C L
- 19. ELEC516/10 Lecture 9 21 2-input NOR Gate Example: Dynamic 2 Input NOR Gate Assume: P(A=1) = 1/2 P(B=1) = 1/2 P(Out=0) = 3/4 Then: CEFF = 3/4 * C L Switching Activity Is Always Higher in Dynamic Circuits
- 20. ELEC516/10 Lecture 9 22 Transition Probabilities for Dynamic Gates Switching Activity for Precharged Dynamic Gates P01 = P0
- 21. ELEC516/10 Lecture 9 24 Glitching in Static CMOS Networks ABC X Z 101 000 Unit Delay A B X Z C • Gates have a nonzero propagation delay resulting in spurious transitions or glitches (dynamic hazards) – glitch: node exhibits multiple transitions in a single cycle before settling to the correct logic value
- 22. ELEC516/10 Lecture 9 25 Example : Adder Circuit 0 5 10 0.0 2.0 4.0 Time, ns Sum Output Voltage, Volts Cin S15 S10 6 5 4 3 2 S1 Add0 Add1 Add2 Add14 Add15 S0 S1 S2 S14 S15 Cin
- 23. ELEC516/10 Lecture 9 26 How to Cope with Glitching? F1 F2 F3 F1 F3 F2 0 0 0 0 1 2 0 0 0 0 1 1 Equalize Lengths of Timing Paths Through Design
- 24. ELEC516/10 Lecture 9 27 Balanced Delay Paths to Reduce Glitching So equalize the lengths of timing paths through logic F1 F2 F3 0 0 0 0 1 2 F1 F2 F3 0 0 0 0 1 1 Glitching is due to a mismatch in the path lengths in the logic network; if all input signals of a gate change simultaneously, no glitching occurs
- 25. ELEC516/10 Lecture 9 28 Glitch Reduction by Pipelining • Glitches depend on the logic depth of the circuit - gates deeper in the logic network are more prone to glitching – arrival times of the gate inputs are more spread due to delay imbalances – usually affected more by primary input switching • Reduce logic depth by adding pipeline registers – additional energy used by the clock and pipeline registers PC Fetch Decode Execute Memory WriteBack Instruction MAR MDR I$ D$ clk pipeline stage isolation register
- 26. ELEC516/10 Lecture 9 29 Short Circuit Power Consumption Finite slope of the input signal causes a direct current path between VDD and GND for a short period of time during switching when both the NMOS and PMOS transistors are conducting. Vin Vout CL Isc
- 27. ELEC516/10 Lecture 9 30 Short Circuit Currents Determinates • Duration and slope of the input signal, tsc • Ipeak determined by – the saturation current of the P and N transistors which depend on their sizes, process technology, temperature, etc. – strong function of the ratio between input and output slopes • a function of CL Esc = tsc VDD Ipeak P01 Psc = tsc VDD Ipeak f01
- 28. ELEC516/10 Lecture 9 31 Impact of CL on Psc Vin Vout CL Isc 0 Vin Vout CL Isc Imax Large capacitive load Output fall time significantly larger than input rise time. Small capacitive load Output fall time substantially smaller than the input rise time.
- 29. ELEC516/10 Lecture 9 32 Ipeak as a Function of CL -0.5 0 0.5 1 1.5 2 2.5 0 2 4 6 time (sec) x 10-10 x 10-4 CL = 20 fF CL = 100 fF CL = 500 fF 500 psec input slope Short circuit dissipation is minimized by matching the rise/fall times of the input and output signals - slope engineering. When load capacitance is small, Ipeak is large.
- 30. ELEC516/10 Lecture 9 33 Static Power Consumption Vin=5V Vout CL Vdd Istat Pstat = P(In=1).Vdd . Istat • Dominates over dynamic consumption • Not a function of switching frequency
- 31. ELEC516/10 Lecture 9 34 Leakage (Static) Power Consumption Sub-threshold current is the dominant factor. All increase exponentially with temperature! VDD Ileakage Vout Drain junction leakage Sub-threshold current Gate leakage
- 32. ELEC516/10 Lecture 9 35 Power consideration – leakage current ) 1 ( / / ) ( kT q V nkT q V V sub ds th gs e e K I K: technology constant; q: electronic charge; k: Boltzman constant N: nonlinearity constant (between 1 and 2); T: Temperature
- 33. ELEC516/10 Lecture 9 36 Leakage as a Function of VT 0 0.2 0.4 0.6 0.8 1 VGS (V) ID (A) VT=0.4V VT=0.1V 10-2 10-12 10-7 Continued scaling of supply voltage and the subsequent scaling of threshold voltage will make subthreshold conduction a dominate component of power dissipation. An 90mV/decade VT roll-off - so each 255mV increase in VT gives 3 orders of magnitude reduction in leakage (but adversely affects performance)
- 34. ELEC516/10 Lecture 9 37 Sub-Threshold in MOS VT=0.6 VT=0.2 ID VGS Lower Bound on Threshold to Prevent Leakage
- 35. ELEC516/10 Lecture 9 38 Low Power Design Space • The dynamic power consumption equation reveals the three degrees of freedom inherent in the low power design space: – Voltage – Physical capacitance – Data activity • Optimization for power entails an attempt to reduce one or more of these factors. Interactions among these factors complicate the optimization problem. • Deep sub-micron design - need to minimize leakage and sub-threshold current
- 36. ELEC516/10 Lecture 9 39 Power Reduction Strategy (I) • Voltage Reduction – 5V ->3.3 V -> 2.5V ->1.8V-> 1.0V – Mixed supplies in system and/or on chip, by using the minimum voltage for different chips of functions within a chip, together with on-chip voltage converters if required. • Low-voltage circuit techniques are required to give good performances even with low voltages • Less noisy structures and better signal integrity handling is required • Lower Vth process is required to maintain good transistor speed performances
- 37. ELEC516/10 Lecture 9 40 Reducing Vdd P x td = Et = CL * Vdd 2 E(Vdd=2) = (CL) * (2)2 (CL) * (5)2 E(Vdd=5) Strong function of voltage (V 2 dependence). Relatively independent of logic function and style. E(Vdd=2) 0.16 E (Vdd =5) 0.03 0.05 0.07 0.1 0.15 0.20 0.30 0.50 0.70 1.00 1.5 1 2 5 51 stage ring oscillator 8-bit adder Vdd (volts) quadratic dependence N O RMALI Z ED PO WER - DELAY P R O DUCT Power Delay Product Improves with lowering V DD .
- 38. ELEC516/10 Lecture 9 41 Lower Vdd Increases Delay C L * Vdd I = Td Td(Vdd=5) Td(Vdd=2) = (2) * (5 - 0.7)2 (5) * (2 - 0.7)2 4 I ~ (Vdd - Vt)2 Relatively independent of logic function and style. 1.00 1.50 2.00 2.50 3.00 3.50 4.00 4.50 5.00 5.50 6.00 6.50 7.00 7.50 2.00 4.00 6.00 Vdd (volts) NORMALIZED DELAY adder (SPICE) microcoded DSP chip multiplier adder ring oscillator clock generator 2.0mm technology
- 39. ELEC516/10 Lecture 9 42 Lowering the Threshold DESIGN FOR PLeakage == PDynamic Vt = 0.2 Vt = 0 I D VGS Reduces the Speed Loss, But Increases Leakage Vdd Delay 2Vt Interesting Design Approach:
- 40. ELEC516/10 Lecture 9 43 What Threshold Voltage to Use? • Energy vs. Vt for a fixed throughput • An “optimal” Vt/Vdd point trades switching and leakage energy
- 41. ELEC516/10 Lecture 9 44 Stack Effect • Leakage is a function of the circuit topology and the value of the inputs VT = VT0 + (|-2fF + VSB| - |-2fF|) where VT0 is the threshold voltage at VSB = 0; VSB is the source- bulk (substrate) voltage; is the body-effect coefficient A B B A Out VX A B VX ISUB 0 0 VT ln(1+n) VGS=VBS= -VX 0 1 0 VGS=VBS=0 1 0 VDD-VT VGS=VBS=0 1 1 0 VSG=VSB=0 • Leakage is least when A = B = 0 • Leakage reduction due to stacked transistors is called the stack effect
- 42. ELEC516/10 Lecture 9 45 Leakage as a Function of Design Time VT • Reducing the VT increases the sub-threshold leakage current (exponentially) – 90mV reduction in VT increases leakage by an order of magnitude • But, reducing VT decreases gate delay (increases performance) 0 0.2 0.4 0.6 0.8 1 VGS (V) ID (A) VT=0.4V VT=0.1V • Determine the critical path(s) at design time and use low VT devices on the transistors on those paths for speed. Use a high VT on the other logic for leakage control. – A careful assignment of VT’s can reduce the leakage by as much as 80%
- 43. ELEC516/10 Lecture 9 46 Dual-Thresholds Inside a Logic Block • Minimum energy consumption is achieved if all logic paths are critical (have the same delay) • Use lower threshold on timing-critical paths – Assignment can be done on a per gate or transistor basis; no clustering of the logic is needed – No level converters are needed
- 44. ELEC516/10 Lecture 9 47 Variable VT (ABB) at Run Time • VT = VT0 + (|-2fF + VSB| - |-2fF|) 0.4 0.45 0.5 0.55 0.6 0.65 0.7 0.75 0.8 0.85 0.9 -2.5 -2 -1.5 -1 -0.5 0 VSB (V) A negative bias on VSB causes VT to increase Adjusting the substrate bias at run time is called adaptive body-biasing (ABB) l Requires a dual well fab process For an n-channel device, the substrate is normally tied to ground (VSB = 0)
- 45. ELEC516/10 Lecture 9 48 Techniques for Burst Mode Computation High VT High VT Low VT SLEEP SLEEP - + - + on standby standby on Vin Vout Vdd Vp >0 Vn < 0 Multiple VT Technology Substrate Bias Controlled Variable VT Devices (Disable high VT devices during idle periods) (increase VT during idle periods) e.g. [Sakata-93], [Mutoh-93] e.g. [Seta-95]
- 46. ELEC516/10 Lecture 9 49 Multiple-Threshold Circuits • Previous approaches cannot reduce leakage power during active mode. • Use dual Vt CMOS logic, low Vt for critical paths and high Vt for non-critical paths, problem - large threshold swing • Triple Vt CMOS circuit to reduce the sub-thresold swing [Fujii et. al. ISSCC-98] – for high speed low power active mode, low and medium Vt are used for critical and non-critical paths, respectively. – For standby mode, high Vt MOSFET is inserted between the supply rail and the virtual supply rail.
- 47. ELEC516/10 Lecture 9 50 Multiple-Threshold Circuits
- 48. ELEC516/10 Lecture 9 51 Power considerations – reduce leakage • Leakage proportional to device width – Use smallest devices for critical path. • Leakage drops with stacked devices (drain voltage divider) – Use stacked transistors for critical path. • Leakage drops with increasing channel length – Slightly increase L for critical Path • Use dual VT process providing two threshold VT – Use high VT transistors for critical path
- 49. ELEC516/10 Lecture 9 52 Power consideration – reduce leakage • Switch off critical path transistors when not needed. • Stand-by mode between supply and virtual supply lines • Stand-by vectors – Apply input vector which minimizes leakage. – Achieved using Mux
- 50. ELEC516/10 Lecture 9 53 Power gating transistor Sizing • The effect of power gating transistor size – As the size decreases, logic performance also decreases. – As the size increases, leakage current and chip area also increase. – Proper sizing is very important. – power gating transistor size should be decided within 2% performance degradation. Vop = VDD - V V must be sized within 2% performance degradation. VDD GND Low Vt HighVt Switch Control
- 51. ELEC516/10 Lecture 9 54 Power Reduction Strategy (II) • Reducing capacitance – Process scaling and better integration, with smaller capacitances in more aggressive processes – Improved devices and interconnect technology – Efficient clock generation and distribution – Good memory hierarchy – In-place optimization using a library containing ranges of gates with different strengths, through replacement of gates to use the optimum drive in the critical paths and minimum drive elsewhere
- 52. ELEC516/10 Lecture 9 55 Reducing Effective Capacitance Global bus architecture Local bus architecture Shared Resources incur Switching Overhead
- 53. ELEC516/10 Lecture 9 56 Power consideration – Reduce Capacitance • Reduce switched capacitance C – Careful transistor sizing, transistor ordering, tighter and more compact layout – Hierarchical architecture and add TG to isolate buses – Segmented structures Shared bus driven by A or B when Sending values to C Insert switch to isolate bus segment when B sending to C
- 54. ELEC516/10 Lecture 9 57 Power Reduction Strategy (III) • Reducing activities – lowering operating frequency – Using power management strategies, such as Gated clocks, Power-Down of non-operational units – Reduce switching activities, Power = Energy/transition * transition rate = • Power dissipation is data dependent and hence is a function of switching activity. • P0->1 is the switching probability • Effective switching capacitance = Ceff=CL* P0->1 f V C f P V C f V C dd eff dd L dd L 2 1 0 2 1 0 2
- 55. ELEC516/10 Lecture 9 58 Factors Affecting Power Consumption - Revisited • Degree of freedom for low power design space: – Voltage – Physical capacitance – Data activity • Power minimization approaches: – Run at minimum allowable voltage – Minimize effective switching capacitances
- 56. ELEC516/10 Lecture 9 59 System Level - Power Down Techniques Operating States: Active or Full-On (fastest clock) Standby (slow clock) Suspend or Sleep (slow clock or shut down) Micro Processor Activity Analyzer
- 57. ELEC516/10 Lecture 9 60 Dynamic Power as a Function of VDD • Decreasing the VDD decreases dynamic energy consumption (quadratically) • But, increases gate delay (decreases performance) 1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 0.8 1 1.2 1.4 1.6 1.8 2 2.2 2.4 VDD (V) • Determine the critical path(s) at design time and use high VDD for the transistors on those paths for speed. Use a lower VDD on the other gates, especially those that drive large capacitances (as this yields the largest energy benefits).
- 58. ELEC516/10 Lecture 9 61 Multiple VDD Considerations • How many VDD? – Two is becoming common – Many chips already have two supplies (one for core and one for I/O) • When combining multiple supplies, level converters are required whenever a module at the lower supply drives a gate at the higher supply (step-up) – If a gate supplied with VDDL drives a gate at VDDH, the PMOS never turns off • The cross-coupled PMOS transistors do the level conversion • The NMOS transistor operate on a reduced supply – Level converters are not needed for a step-down change in voltage – Overhead of level converters can be mitigated by doing conversions at register boundaries and embedding the level conversion inside the flipflop VDDH Vin Vout VDDL
- 59. ELEC516/10 Lecture 9 62 Dual-Supply Inside a Logic Block • Minimum energy consumption is achieved if all logic paths are critical (have the same delay) • Clustered voltage-scaling – Each path starts with VDDH and switches to VDDL (gray logic gates) when delay slack is available – Level conversion is done in the flipflops at the end of the paths
- 60. ELEC516/10 Lecture 9 63 Power Conscious Behavioral Design • Run functional units at the minimum allowed voltage while satisfying the timing constraints • Parallelize or pipeline data-path, memory and controllers to compensate for throughput loss due to reduced supply voltage • Power down functional units which are not is use; Put in “dynamic power management” capability • Avoid centralized resources (controllers, functional blocks, global busses, etc.) as much possible • Map functions to hardware so that inter-chip communication is minimized • Schedule and bind operations to functional units so as to reduce the activity of the input operands • Reorder operands to reduce switching activity; Keep the inputs to an idle unit unchanged
- 61. ELEC516/10 Lecture 9 64 Low Power Datapath Design - reducing the supply voltage • Reducing supply voltage has quadratic effect on power saving, but a negative effect on performance. • Performance can be gained back by logical and architectural optimizations, e.g. lookahead adder instead of ripple-carry adder, using parallelism to increase performance
- 62. ELEC516/10 Lecture 9 65 Power Considerations – reduce f and VDD • Reducing frequency does not save energy, just reduces rate at which it is consumed – Power is lower but system must run longer • Reducing supply voltage is very effective (reduce voltage by 0.5 improves energy/transition by 0.25). • Dropping the voltage will result in reduced performance in terms of speed (need to recover the performance using parallelism). • Trade surplus performance for lower energy by reducing the supply voltage until performance are as required
- 63. ELEC516/10 Lecture 9 66 Power considerations – Dynamic voltage Scaling Can run at lower voltage and hence improves quadraticaly power comsp. – 8-bit adder/comparator: (Chandrakasan er. Al.) • consumes Pref at 40Mhz at 5V with Area= 0.53mm2 – Two parallel interleaved architecture: • Consumes 0.36Pref at 20MHz, 2.9V with Area=1.80mm2 – One Pipelined architecture: • Consumes 0.39Pref at 40MHz, 2.9V with Area=0.69mm2 – Pipelined and parallel • Consumes 0.2Pref at 20MHz, 2.0V with Area=1.96mm2
- 64. ELEC516/10 Lecture 9 67 Minimizing the power consumption using parallelism • Reference Design Critical path = 25ns, clock frequency = 40MHz Supply voltage = 5V ref ref ref ref f V C P 2
- 65. ELEC516/10 Lecture 9 68 • Parallel implementation of the reference design New critical path = 50nsec, Cpar->2.15 Cref, Vpar ->2.9V, fpar ->0.5 fref ref ref ref ref par par par par P f V C f V C P 36 . 0 ) 2 ( ) 58 . 0 )( 15 . 2 ( 2 2
- 66. ELEC516/10 Lecture 9 69 Pipelined Data Path • Critical path delay is less => max[Tadder, Tcomparator]. • Keeping clock rate constant: fpipe=fref, Voltage can be dropped to Vpipe = Vref/1.7, while maintaining the original througput • Capacitance slightly higher: Cpipe=1.15Cref • Ppipe=(1.15Cref)(Vref/1.7)2fref » 0.39Pref
- 67. ELEC516/10 Lecture 9 70 How low a voltage can be used • Capacitance overhead starts to dominate at “high” levels of parallelism and results in an optimum voltage.
- 68. ELEC516/10 Lecture 9 71 Voltage as a design variable Adapting Voltage to Workload yields cubic reduction
- 69. ELEC516/10 Lecture 9 72 Multiple supply voltages – Filter Example 1 2 3 4 5 6 7 8 9 10 * * * * * * * * + + * * * * + + + + + + + + + + + + + + Power (5V)/Power(5V,3V,2.4V) = 1.5 [Raje95] 2.4V 3V 5V
- 70. ELEC516/10 Lecture 9 73 Using Multiple Voltages C1 C2 C3 Vdd1 Vdd2 Vdd3 Vdd1Vdd2 Vdd3 critical path non-critical Vdd1 Vdd2 Vdd1Vdd2 [Horowitz-95]
- 71. ELEC516/10 Lecture 9 74 Processor Usage Model • System Optimizations: – Maximize Peak Throughput – Minimize Average Energy/operation – Maximize computation per battery life Desired Throughput time Compute-intensive and Low-latency processes Ceiling: set by top Speed of the processor Single-user system not always computing Background and high- latency processes
- 72. ELEC516/10 Lecture 9 75 Scale Supply Voltage with fclk
- 73. ELEC516/10 Lecture 9 76 Dynamic Voltage Scaling Implementation • VCO: ring oscillator which matches mP critical path • DC-DC: perform D/A, converts battery to regulated Vdd • Provides both voltage regulation and clock generation. Loop Filter DC-DC VCO mP - + Mfref fout Frequency detector battery Vdd
- 74. ELEC516/10 Lecture 9 77 Dynamic Voltage Scaling in Practice Fixed Throughput, Energy/operation Occasionally Demand Peak Throughput Throughput = 8MIPS Energy/ops = 0.24nJ/inst. 1.92mW Throughput = 100MIPS Energy/ops = 2.2nJ/inst. 220mW Peak Throughput = 100MIPS Average Energy/op. = 0.24nJ/inst (~1.8mW) DVS only advantageous when a majority of computation is performed at low throughput
- 75. ELEC516/10 Lecture 9 78 Low-voltage Switching Regulator • Arbitrary Vdd (<Vin) generated using the Buck converter – Vdd = Vin Duty Cycle at Node X • Chief sources of inefficiencies: – Conduction loss (I2R) – Switching loss (CxVin 2fs and LsI2fs) – Gate-drive loss (CgVin 2fs) Page 17 [Stratakos-94]
- 76. ELEC516/10 Lecture 9 79 Adaptive Power Supply Voltage • Exploit Data Dependent Computation Times To vary the Supply [Nielsen94] REG FIFO FIFO REG Control Self-timed Processor Power Supply Vdd(t)
- 77. ELEC516/10 Lecture 9 80 Variable Supply Voltage Control Scheme
- 78. ELEC516/10 Lecture 9 81 Voltage Scheduling for variable workload system • Voltage scheduling under timing constraints • Example [Ishihara-98] – Energy consumption of a processor: • 10nJ/cycle at 2.5V • 25nJ/cycle at 4 V • 40nJ/cycle at 5V – maximum clock frequencies: • 50MHz at 5V, 40MHz at 4V, 25MHz at 2.5V – Given that an application needs 1000M cycles to finish and the timing constraint is 25sec.
- 79. ELEC516/10 Lecture 9 82 Different Voltage Schedules 0 5 10 15 20 25 Time(sec) 5.02 1000Mcycles 50MHz 40J (A) 0 5 10 15 20 25 Time(sec) 5.02 750Mcycles 50MHz 32.5J (B) 0 5 10 15 20 25 Time(sec) 5.02 1000Mcycles 40MHz 25J (C) Timing constraint 2.52 250Mcycles 25MHz 4.02 Energy consumption ( V dd 2 )
- 80. ELEC516/10 Lecture 9 83 Reduce Power Further by Buffering Two samples processed every two sample periods -> increased latency
- 81. ELEC516/10 Lecture 9 84 Example of Buffering Block 1 Block 2 Block 3 Block 4 Block 1, 2 Block 1, 2 Block 3,4 Block 3,4 Tsample Tsample Vdd =5 Vdd =2.5 Vdd =5 Vdd =2.5 Vdd =3.75 Vdd =3.75 Vdd =3.75 Vdd =3.75 C C C Energysample 14 2 ) 5 . 2 ( 2 1 ) 5 ( 2 2 C C Energysample 5 . 10 2 ) 75 . 3 ( ) 75 . 0 ( 2 2
- 82. ELEC516/10 Lecture 9 85 Voltage Island Concept Trade off power for delay by running functional blocks at different voltages Can use mix of Low and High Vt to balance performance and leakage Switch off inactive blocks to reduce leakage power Requires IP standards for power management, clock gating, etc. Delay vs. Voltage 30 25 20 15 10 5 0 Ddelay (ps) 0.7 0.8 0 .9 1.0 1.1 1.2 1.3 Voltage (Vdd) Std. Vt Low Vt E.g.: Telecom ASIC with 1.0/1.2 V islands saved : 16 % active power 50 % standby power Power Management Unit SWITCH SWITCH Logic Low VT Logic Vddo Vdd1 Vdd2 IP1 IP2 Source from Bergamaschi *Slide from Prof Kyung of KAIST
- 83. ELEC516/10 Lecture 9 86 Power Management I/O’s, VReg, Gnd Memory Arrays Vdd 4 High Vt device arrays Optimized for low active power Memory Arrays Vdd 3 Low Vt device arrays Optimized for low active power Microcontroller Vdd 2 DSP Vdd 2 ROM Vdd 1 Monitor Logic Vdd 4 ROM Vdd1 RLM 1 RLM 2 Memory Arrays Vdd 3 Low Vt device arrays Optimized for low active power I/O’s, VReg, Gnd Analog Vdd 5 RLM 3 Vdd 1 I/O’s, VReg, Gnd I/O’s, VReg, Gnd Independently controlled domain power switches Multiple On-Chip Voltage Islands On-Chip Voltage Regulators *Slide from Prof Kyung of KAIST
- 84. ELEC516/10 Lecture 9 87 Controlling VDD and VTH for low power Active Stand-by Multiple VTH Dual-VTH MTCMOS Variable VTH VTH hopping VTCMOS Multiple VDD Dual-VDD Boosted gate MOS Variable VDD VDD hopping Software-hardware cooperation Technology-circuit cooperation MTCMOS : Multi-Threshold CMOS VTCMOS : Variable Threshold CMOS Multiple : spatial assignment Variable : temporal assignment *Slide from Prof Kyung of KAIST
- 85. ELEC516/10 Lecture 9 88 RTL-level optimization-Reducing effective switching activity • General Principle: Avoid Waste. – Application-specific processing – Resource sharing/Locality of reference – Data representations – Preservation of Data correlations – architectural restructuring – Distributed processing – Demand-driven/data-driven computation
- 86. ELEC516/10 Lecture 9 89 Application Specific Processing Application Specific Processing Reduces “Implementation Overhead”
- 87. ELEC516/10 Lecture 9 90 Eliminating Redundant Computation • Dynamically vary the number of operations per sample. • Trade power consumption and filter quality [Ludwig- 96] fs fs fs fs fs …. Out Power Down
- 88. ELEC516/10 Lecture 9 91 Eliminating Redundant Computation Switched Capacitance Reduction ~= Peak Number of Operations Average Number of Operations Strong Function of Signal Statistics ~=2
- 89. ELEC516/10 Lecture 9 92 Reducing switching activity • Multiplexing multiple operations on a single hardware unit can have detrimental effect on the power consumption, because the switching activity may be increased, e.g. shared bus
- 90. ELEC516/10 Lecture 9 93 Bus Multiplexing • Share long data buses with time multiplexing (S1 uses even cycles, S2 odd) S2 S1 D1 D2 S1 S2 D2 D1 • Buses are a significant source of power dissipation due to high switching activities and large capacitive loading – 15% of total power in Alpha 21064 – 30% of total power in Intel 80386 • But what if data samples are correlated (e.g., sign bits)?
- 91. ELEC516/10 Lecture 9 94 Reducing the Effective Capacitance • Circuit and Logic Style - select a circuit style with low capacitance and/or switching activity, e.g. 8-bit adder
- 92. ELEC516/10 Lecture 9 95 Reducing Glitching Activity • Some circuit structures can be the cause of spurious transients, e.g. a 16-bit ripple carry adder • Glitches can be reduced by selecting structures that have balanced signal paths, e.g. tree. • The Brent-Kung lookahead adder and Wallace tree multiplier both have this properties, thus more power attractive.
- 93. ELEC516/10 Lecture 9 96 Data Representation Two’s complement Sign Magnitude • Sign-extension activity significantly reduced using sign magnitude representation. • An accumulator example: sign magnitude datapath switches 30% less capacitance for uniformly distributed inputs
- 94. ELEC516/10 Lecture 9 97 Data Representation – Accumulator Example Sign magnitude datapath switches 30% less capacitance for uniformly distributed inputs
- 95. ELEC516/10 Lecture 9 98 Two’s Complement vs. Sign-Magnitude Two’s complement datapath has a significantly higher switching activity
- 96. ELEC516/10 Lecture 9 99 Bus encoding to reduce switching activity • Minimizing temporal bit transition activity by data representation • Bit encoding: – Active high encoding • high-level voltage for 1, low-level voltage for 0 – Transition-based encoding • voltage change identifies logic 1, no voltage range identifies logic 0 • Word encoding – assign patterns of 1’s and 0’s to each word of information – Non-redundant codes vs. redundant codes • Example of low power coding – Limited-weight code – Gray code – One-hot code – Bus-invert code
- 97. ELEC516/10 Lecture 9 100 Example of Bit Encoding • Reducing average no. of switching on a data bus • transition-based encoding may limit the no. of transitions for non equiprobable input lines • Let p(0) > p(1) and no temporal correlation exists. If active-high encoding is used, the average no. of transition is • For transition-based encoding, it is simply p(1). If p(1) << p(0), transition-based is better than active-high encoding. • To reduce transition, the input patterns are transformed in such a way that the p(0) and p(1) prob. of each input line becomes as different as possible, and then to apply a transition-based encoding before the data is transmitted. ) 1 ( )) 1 ( 1 ( 2 ) 0 ( ) 1 ( ) 1 ( ) 0 ( p p p p p p
- 98. ELEC516/10 Lecture 9 101 Word encoding • One-hot coding •Gray coding - good for sequential data, e.g. addressing for microprocessor [Su-94] •Disadvantage of Gray code - only good for address bus, not for data bus, additional conversion circuitry is needed
- 99. ELEC516/10 Lecture 9 102 Word encoding (cont.) • Bus-invert encoding - use redundancy to save power. • Given a n bit data bus with 2n patterns to be represented and assumes all the patterns are equiprobable and no temporal correlation. p(0) = p(1) = 0.5, the average no. of transition is n/2 per cycle, while the worst case transition is n. • Bus-invert coding - add an extra line to the bus, I, and then comparing the consecutive patterns before transmission. Tw cases – - If the Hamming distance between the two patterns =< n/2, the current pattern is transmitted as it is and I is set to 0. – - if the Hamming distance between the two patterns > n/2, the current pattern is first inverted and then transmitted, and I is se to 1. • Max. transition is limited to n/2 and average transition is reduced by 25% • Drawback - extra line I to indicate whether a pattern has been inverted.
- 100. ELEC516/10 Lecture 9 103 Conditional Inversion Coding
- 101. ELEC516/10 Lecture 9 104 Bus encoding for cross coupling cap. • Cross-coupling cap dominates for very deep sub- micron technology, e.g. < 70nm • Bus model – Stand alone cap. Cs – Cross coupling cap. Cc • Stand alone switching – Apply to single bit line – 0-1 transition • Cross coupling switching – Occurs on adjacent wires – Four types of coupling transition H --> L H --> L L --> H L --> L H --> L L --> H H --> L H --> L Type 1 Type 2 Type 3 Type 4 0 1 2 0 Cs Cc Cc Cs Cs bit 0 bit 1 bit 2
- 102. ELEC516/10 Lecture 9 105 Permutation Based Address code • Rearrange the physical order of the bit lines • Work efficiently on address bus (40%), but not on instruction bus (4%) – Correlation is lower – Permutation is fixed Sender Receiver
- 103. ELEC516/10 Lecture 9 106 Dynamic Reconfigurable Bus Encoding Scheme for Instruction Bus
- 104. ELEC516/10 Lecture 9 107 Overview of the Scheme Decoding information stored as header Decoding information are loaded to LUT first Instruction is called Decoding information stored in LUT will control the Cross_bar Instructions enter Cross_bar to decode Encoding . . . MEM B1 Bn Mem_bus Processing Element . . B1 Bm CACHE CPU Core Cache_bus Decoder (Cross_bar) Look-Up Table Address_bus Target Processor Decoding Computer Target Processor Bit lines reordering Encoding during compilation Mem
- 105. ELEC516/10 Lecture 9 108 Demand/Data-driven operation • Clocking strategy – Gated clocking • System Power Down • Computing Paradigm
- 106. ELEC516/10 Lecture 9 109 Logic Level Power Down Technique • Activity driven - precomputation-based sequential logic optimization – Selectively precompute the output logic values of the circuit one clock cycle before they are required, and use the precomputed values to reduce internal switching activity in the succeeding cycle It is required that g1 = 1 -> f = 1 g2 = 1 -> f = 0 A g2 LE Original Circuit Modified Circuit f A R1 R1 R2 FF FF g1 R2
- 107. ELEC516/10 Lecture 9 110 An example: n-bit comparator • This circuit compares two n-bit numbers A & and computes the function A > B • In general, precomputation works best when there are a small number of complex functions corresponding to the logic block A

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